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rev. 0 information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a adp1111 one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 617/329-4700 world wide web site: http://www.analog.com fax: 617/326-8703 ? analog devices, inc., 1996 micropower, step-up/step-down sw regulator; adjustable and fixed 3.3 v, 5 v, 12 v functional block diagrams driver i lim sw1 sw2 v in gnd set a0 gain block/ error amp comparator a1 a2 fb 1.25v reference oscillator adp1111 driver i lim sw1 sw2 v in gnd set a0 gain block/ error amp comparator a1 a2 sense 1.25v reference oscillator adp1111-5 adp1111-12 r1 r2 220k features operates from 2 v to 30 v input voltage range 72 khz frequency operation utilizes surface mount inductors very few external components required operates in step-up/step-down or inverting mode low battery detector user adjustable current limit internal 1 a power switch fixed or adjustable output voltage 8-pin dip or so-8 package applications 3 v to 5 v, 5 v to 12 v step-up converters 9 v to 5 v, 12 v to 5 v step-down converters laptop and palmtop computers cellular telephones flash memory vpp generators remote controls peripherals and add-on cards battery backup supplies uninterruptible supplies portable instruments general description the adp1111 is part of a family of step-up/step-down switch- ing regulators that operates from an input voltage supply of 2 v to 12 v in step-up mode and up to 30 v in step-down mode. the adp1111 can be programmed to operate in step-up/step- down or inverting applications with only 3 external components. the fixed outputs are 3.3 v, 5 v and 12 v; and an adjustable version is also available. the adp1111 can deliver 100 ma at 5 v from a 3 v input in step-up mode, or it can deliver 200 ma at 5 v from a 12 v input in step-down mode. maximum switch current can be programmed with a single resistor, and an open collector gain block can be arranged in multiple configuration for low battery detection, as a post linear regulator, undervoltage lockout, or as an error amplifier. if input voltages are lower than 2 v, see the adp1110. rev. a one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ?2009 analog devices, inc. all rights reserved. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ?1996C2009 analog devices, inc. all rights reserved.
C2C rev. 0 adp1111Cspecifications parameter conditions v s min typ max units quiescent current switch off i q 300 500 a input voltage step-up mode v in 2.0 12.6 v step-down mode 30.0 v comparator trip point voltage adp1111 1 1.20 1.25 1.30 v output sense voltage adp1111-3.3 v out 3.13 3.30 3.47 v adp1111-5 2 4.75 5.00 5.25 v adp1111-12 2 11.40 12.00 12.60 v comparator hysteresis adp1111 8 12.5 mv output hysteresis adp1111-3.3 21 50 mv adp1111-5 32 50 mv adp1111-12 75 120 mv oscillator frequency f osc 54 72 88 khz duty cycle full load dc 43 50 65 % switch on time i lim tied to v in t on 57 9 s sw saturation voltage t a = +25 c step-up mode v in = 3.0 v, i sw = 650 ma v sat 0.5 0.65 v v in = 5.0 v, i sw = 1 a 0.8 1.0 v step-down mode v in = 12 v, i sw = 650 ma 1.1 1.5 v feedback pin bias current adp1111 v fb = 0 v i fb 160 300 na set pin bias current v set = v ref i set 270 400 na gain block output low i sink = 300 a v set = 1.00 v v ol 0.15 0.4 v reference line regulation 5 v v in 30 v 0.02 0.075 %/v 2 v v in 5 v 0.4 %/v gain block gain r l = 100 k 3 a v 1000 6000 v/v current limit t a = +25 c 220 from i lim to v in i lim 400 ma current limit temperature coefficient C0.3 %/ c switch off leakage current t a = +25 c measured at sw1 pin v sw1 = 12 v 1 10 a maximum excursion below gnd t a = +25 c i sw1 10 a, switch off C400 C350 mv notes 1 this specification guarantees that both the high and low trip points of the comparator fall within the 1.20 v to 1.30 v range. 2 the output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. the output voltage on the fixed out put versions will always be within the specified range. 3 100 k resistor connected between a 5 v source and the ao pin. 4 all limits at temperature extremes are guaranteed via correlation using standard statistical methods. specifications subject to change without notice. (0 c t a +70 c, v in = 3 v unless otherwise noted) rev. a
adp1111 rev.a 3 absolute maximum ratings parameter rating supply voltage 36 v sw1 pin voltage 50 v sw2 pin voltage ?0.5 v to v in feedback pin voltage (adp1111) 5.5 v switch current 1.5 a maximum power dissipation 500 mw operating temperature range adp1111a 0c to 70c storage temperature range ?65c to +150c lead temperature (soldering, 10 sec) 300c typical application 10f (optional) i lim v in sw1 sw2 gnd adp1111ar-5 sumid a cd54-220k 22h 5v 100ma 33f sense 3v input mbrs120t3 figure 1. 3 v to 5 v step-up converter esd caution pin descriptions mnemonic function i lim for normal conditions this pin is connected to v in . when lower current is required, a resistor should be connected between i lim and v in . limiting the switch current to 400 ma is achieved by connecting a 220 resistor. v in input voltage. sw1 collector node of power transistor. for step-down con-figuration, connect to v in . for step-up configu- ration, connect to an inductor/diode. sw2 emitter node of power transistor. for step-down configuration, connect to inductor/diode. for step-up configuration, connect to ground. do not allow this pin to go more than a diode drop below ground. gnd ground. ao auxiliary gain (gb) output. the open collector can sink 300 a. it can be left open if unused. set gain amplifier input. the amplifiers positive input is connected to set pin and its negative input is con- nected to the 1.25 v reference. it can be left open if unused. fb/sense on the adp1111 (adjustable) version this pin is con- nected to the comparator input. on the adp1111-3.3, adp1111-5 and adp1111-12, the pin goes directly to the internal application resistor that sets output voltage. pin configurations a0 i lim s w1 gnd v in s w2 *fixed versions fb (sense)* set 1 2 3 4 8 7 6 5 adp1111 top view (not to scale) 8-lead plastic dip (n-8) a0 i lim sw1 gnd v in sw2 *fixed versions fb (sense) * set 1 2 3 4 8 7 6 5 adp1111 top view (not to scale) 8-lead plastic soic (so-8)
i switch current ? a 1.4 0 0.1 0.2 0.4 0.6 0.8 1.0 1.2 1.2 0.6 0.4 0.2 1.0 0.8 v in = 2v v in = 3v v in = 5v saturation voltage ? v figure 2. saturation voltage vs. i switch current in step-up mode i switch current ? a 2.0 0 0.1 0.2 0.4 0.6 0.8 0.9 1.8 0.6 0.4 0.2 1.6 1.4 v in = 12v on voltage ? v 1.2 1.0 0.8 figure 3. switch on voltage vs. i switch current in step-down mode input voltage ? v 1.5 30 3 6 9 12 15 18 21 24 27 0 1400 1200 1000 800 600 400 200 quiescent current quiescent current ? a figure 4. quiescent current vs. input voltage adp1111Ctypical characteristics C4C rev. 0 76 71 67 230 4 6 8 1012151821 2427 75 72 70 69 74 73 input voltage ? v oscillator frequency ? khz oscillator frequency 68 figure 5. oscillator frequency vs. input voltage r lim ? 1.9 1.7 0.1 1 1000 10 100 1.5 1.3 0.5 1.1 0.9 0.7 0.3 switch current ? a step-down with v in = 12v step-up with 2v < v in < 5v figure 6. maximum switch current vs. r lim oscillator frequency ? khz temperature ? c 80 70 60 ?40 85 25 64 62 68 66 oscillator frequency 70 0 78 72 76 74 figure 7. oscillator frequency vs. temperature rev. a
adp1111 C5C rev. 0 temperature ? c 7.5 6.6 ?40 85 25 7.3 7.2 6.8 6.7 7.4 on time on time ? s 7.1 6.9 7.0 070 figure 8. switch on time vs. temperature temperature ? c 58 56 46 ?40 85 25 54 48 duty cycle duty cycle ? % 52 50 070 figure 9. duty cycle vs. temperature temperature ? c 0.6 0.5 0 ?40 85 25 0.3 0.2 0.1 0.4 v in = 3v @ i sw = 0.65a saturation voltage ? v 070 figure 10. saturation voltage vs. temperature in step-up mode temperature ? c 1.10 1.05 0.80 ?40 85 25 1.00 0.85 v in = 12v @ i sw = 0.65a on voltage ? v 0.95 0.90 070 figure 11. switch on voltage vs. temperature in step- down mode temperature ? c 500 0 ?40 85 25 300 250 200 100 400 350 quiescent current quiescent current ? a 150 50 070 450 figure 12. quiescent current vs. temperature temperature ? c 250 0 ?40 85 25 150 100 50 200 bias current bias current ? a 070 figure 13. feedback bias current vs. temperature rev. a
adp1111 C6C rev. 0 theory of operation the adp1111 is a flexible, low-power, switch-mode power supply (smps) controller. the regulated output voltage can be greater than the input voltage (boost or step-up mode) or less than the input (buck or step-down mode). this device uses a gated-oscillator technique to provide very high performance with low quiescent current. a functional block diagram of the adp1111 is shown on the first page of this data sheet. the internal 1.25 v reference is connected to one input of the comparator, while the other input is externally connected (via the fb pin) to a feedback network connected to the regulated output. when the voltage at the fb pin falls below 1.25 v, the 72 khz oscillator turns on. a driver amplifier provides base drive to the internal power switch, and the switching action raises the output voltage. when the voltage at the fb pin exceeds 1.25 v, the oscillator is shut off. while the oscillator is off, the adp1111 quiescent current is only 300 a. the comparator includes a small amount of hysteresis, which ensures loop stability without requiring external compo- nents for frequency compensation. the maximum current in the internal power switch can be set by connecting a resistor between v in and the i lim pin. when the maximum current is exceeded, the switch is turned off. the current limit circuitry has a time delay of about 1 s. if an external resistor is not used, connect i lim to v in . further information on i lim is included in the applications section of this data sheet. the adp1111 internal oscillator provides 7 s on and 7 s off times that are ideal for applications where the ratio between v in and v out is roughly a factor of two (such as converting +3 v to + 5 v). however, wider range conversions (such as generating +12 v from a +5 v supply) can easily be accomplished. an uncommitted gain block on the adp1111 can be connected as a low-battery detector. the inverting input of the gain block is internally connected to the 1.25 v reference. the noninverting input is available at the set pin. a resistor divider, connected between v in and gnd with the junction connected to the set pin, causes the ao output to go low when the low battery set point is exceeded. the ao output is an open collector npn transistor that can sink 300 a. the adp1111 provides external connections for both the collector and emitter of its internal power switch that permit both step-up and step-down modes of operation. for the step- up mode, the emitter (pin sw2) is connected to gnd, and the collector (pin sw1) drives the inductor. for step-down mode, the emitter drives the inductor while the collector is connected to v in . the output voltage of the adp1111 is set with two external resistors. three fixed-voltage models are also available: adp1111C3.3 (+3.3 v), adp1111C5 (+5 v) and adp1111C12 (+12 v). the fixed-voltage models are identical to the adp1111, except that laser-trimmed voltage-setting resistors are included on the chip. on the fixed-voltage models of the adp1111, simply connect the feedback pin (pin 8) directly to the output voltage. component selection general notes on inductor selection when the adp1111 internal power switch turns on, current begins to flow in the inductor. energy is stored in the inductor core while the switch is on, and this stored energy is transferred to the load when the switch turns off. since both the collector and the emitter of the switch transistor are accessible on the adp1111, the output voltage can be higher, lower, or of opposite polarity than the input voltage. to specify an inductor for the adp1111, the proper values of inductance, saturation current and dc resistance must be determined. this process is not difficult, and specific equations for each circuit configuration are provided in this data sheet. in general terms, however, the inductance value must be low enough to store the required amount of energy (when both input voltage and switch on time are at a minimum) but high enough that the inductor will not saturate when both v in and switch on time are at their maximum values. the inductor must also store enough energy to supply the load, without saturating. finally, the dc resistance of the inductor should be low so that excessive power will not be wasted by heating the windings. for most adp1111 applications, an inductor of 15 h to 100 h with a saturation current rating of 300 ma to 1 a and dc resistance <0.4 is suitable. ferrite-core inductors that meet these specifications are available in small, surface- mount packages. to minimize electro-magnetic interference (emi), a toroid or pot-core type inductor is recommended. rod-core inductors are a lower-cost alternative if emi is not a problem. calculating the inductor value selecting the proper inductor value is a simple three step process: 1. define the operating parameters: minimum input voltage, maximum input voltage, output voltage and output current. 2. select the appropriate conversion topology (step-up, step- down, or inverting). 3. calculate the inductor value using the equations in the following sections. temperature ? c 350 300 0 ?40 85 25 200 150 100 50 250 bias current bias current ? a 070 figure 14. set pin bias current vs. temperature rev. a
adp1111 C7C rev. 0 inductor selectionCstep-up converter in a step-up or boost converter (figure 18), the inductor must store enough power to make up the difference between the input voltage and the output voltage. the power that must be stored is calculated from the equation: p l = v out + v d ? v in ( min ) () ? i out () (equation 1) where v d is the diode forward voltage (0.5 v for a 1n5818 schottky). because energy is only stored in the inductor while the adp1111 switch is on, the energy stored in the inductor on each switching cycle must be equal to or greater than: p f l osc (equation 2) in order for the adp1111 to regulate the output voltage. when the internal power switch turns on, current flow in the inductor increases at the rate of: i l t () = v in r ' 1 ? e ? r ' t l ? ? ? ? ? ? (equation 3) where l is in henrys and r' is the sum of the switch equivalent resistance (typically 0.8 at +25 c) and the dc resistance of the inductor. in most applications, the voltage drop across the switch is small compared to v in so a simpler equation can be used: i l t () = v in l t (equation 4) replacing t in the above equation with the on time of the adp1111 (7 s, typical) will define the peak current for a given inductor value and input voltage. at this point, the inductor energy can be calculated as follows: e l = 1 2 l ? i 2 peak (equation 5) as previously mentioned, e l must be greater than p l /f osc so that the adp1111 can deliver the necessary power to the load. for best efficiency, peak current should be limited to 1 a or less. higher switch currents will reduce efficiency because of increased saturation voltage in the switch. high peak current also increases output ripple. as a general rule, keep peak current as low as possible to minimize losses in the switch, inductor and diode. in practice, the inductor value is easily selected using the equations above. for example, consider a supply that will generate 12 v at 40 ma from a 9 v battery, assuming a 6 v end-of-life voltage. the inductor power required is, from equation 1: p l = 12 v + 0.5 v ? 6 v () ? 40 ma () = 260 mw on each switching cycle, the inductor must supply: p l f osc = 260 mw 72 khz = 3.6 j since the required inductor power is fairly low in this example, the peak current can also be low. assuming a peak current of 500 ma as a starting point, equation 4 can be rearranged to recommend an inductor value: l = v in i l ( max ) t = 6 v 500 ma 7 s = 84 h substituting a standard inductor value of 68 h with 0.2 dc resistance will produce a peak switch current of: i peak = 6 v 1. 0 1 ? e ? 1. 0 ? 7 s 68 h ? ? ? ? ? ? = 587 ma once the peak current is known, the inductor energy can be calculated from equation 5: e l = 1 2 68 h () ? 587 ma () 2 = 11.7 j since the inductor energy of 11.7 j is greater than the p l /f osc requirement of 3.6 j , the 68 h inductor will work in this application. by substituting other inductor values into the same equations, the optimum inductor value can be selected. when selecting an inductor, the peak current must not exceed the maximum switch current of 1.5 a. if the equations shown above result in peak currents > 1.5 a, the adp1110 should be considered. since this device has a 70% duty cycle, more energy is stored in the inductor on each cycle. this results is greater output power. the peak current must be evaluated for both minimum and maximum values of input voltage. if the switch current is high when v in is at its minimum, the 1.5 a limit may be exceeded at the maximum value of v in . in this case, the adp1111s current limit feature can be used to limit switch current. simply select a resistor (using figure 6) that will limit the maximum switch current to the i peak value calculated for the minimum value of v in . this will improve efficiency by producing a constant i peak as v in increases. see the limiting the switch current section of this data sheet for more information. note that the switch current limit feature does not protect the circuit if the output is shorted to ground. in this case, current is only limited by the dc resistance of the inductor and the forward voltage of the diode. inductor selectionCstep-down converter the step-down mode of operation is shown in figure 19. unlike the step-up mode, the adp1111s power switch does not saturate when operating in the step-down mode; therefore, switch current should be limited to 650 ma in this mode. if the input voltage will vary over a wide range, the i lim pin can be used to limit the maximum switch current. higher switch current is possible by adding an external switching transistor as shown in figure 21. the first step in selecting the step-down inductor is to calculate the peak switch current as follows: i peak = 2 i out dc v out + v d v in ? v sw + v d ? ? ? ? ? ? (equation 6) where dc = duty cycle (0.5 for the adp1111) v sw = voltage drop across the switch v d = diode drop (0.5 v for a 1n5818) i out = output current v out = the output voltage v in = the minimum input voltage rev. a
adp1111 C8C rev. 0 as previously mentioned, the switch voltage is higher in step- down mode than in step-up mode. v sw is a function of switch current and is therefore a function of v in , l, time and v out . for most applications, a v sw value of 1.5 v is recommended. the inductor value can now be calculated: l = v in min () ? v sw ? v out i peak ? t on (equation 7) where t on = switch on time (7 s). if the input voltage will vary (such as an application that must operate from a 9 v, 12 v or 15 v source), an r lim resistor should be selected from figure 6. the r lim resistor will keep switch current constant as the input voltage rises. note that there are separate r lim values for step-up and step-down modes of operation. for example, assume that +5 v at 300 ma is required from a +12 v to +24 v source. deriving the peak current from equation 6 yields: i peak = 2 ? 300 ma 0.5 5 + 0.5 12 ? 1. 5 + 0.5 ? ? ? ? ? ? = 600 ma then, the peak current can be inserted into equation 7 to calculate the inductor value: l = 12 ? 1. 5 ? 5 600 ma ? 7 s = 64 h since 64 h is not a standard value, the next lower standard value of 56 h would be specified. to avoid exceeding the maximum switch current when the input voltage is at +24 v, an r lim resistor should be specified. using the step-down curve of figure 6, a value of 560 will limit the switch current to 600 ma. inductor selectionCpositive-to-negative converter the configuration for a positive-to-negative converter using the adp1111 is shown in figure 22. as with the step-up converter, all of the output power for the inverting circuit must be supplied by the inductor. the required inductor power is derived from the formula: p = i l out vv out d + () ? () (equation 8) the adp1111 power switch does not saturate in positive-to- negative mode. the voltage drop across the switch can be modeled as a 0.75 v base-emitter diode in series with a 0.65 resistor. when the switch turns on, inductor current will rise at a rate determined by: i l t () = v l r ' 1 ? e ? r ' t l ? ? ? ? ? ? (equation 9) where: r' = 0.65 + r l(dc) v l = v in C 0.75 v for example, assume that a C5 v output at 50 ma is to be generated from a +4.5 v to +5.5 v source. the power in the inductor is calculated from equation 8: p l = | ? 5 v | + 0.5 v | () ? 50 ma () = 275 mw during each switching cycle, the inductor must supply the following energy: p l f osc = 275 mw 72 khz = 3.8 j using a standard inductor value of 56 h with 0.2 dc resistance will produce a peak switch current of: i peak = 4.5 v ? 0.75 v 0.65 + 0.2 1 ? e ? 0.85 ? 7 s 56 h ? ? ? ? ? ? = 445 ma once the peak current is known, the inductor energy can be calculated from (equation 9): e l = 1 2 56 h () ? 445 ma () 2 = 5.54 j since the inductor energy of 5.54 j is greater than the p l /f osc requirement of 3.82 j , the 56 h inductor will work in this application. the input voltage only varies between 4.5 v and 5.5 v in this application. therefore, the peak current will not change enough to require an r lim resistor and the i lim pin can be connected directly to v in . care should be taken, of course, to ensure that the peak current does not exceed 650 ma. capacitor selection for optimum performance, the adp1111s output capacitor must be selected carefully. choosing an inappropriate capacitor can result in low efficiency and/or high output ripple. ordinary aluminum electrolytic capacitors are inexpensive but often have poor equivalent series resistance (esr) and equivalent series inductance (esl). low esr aluminum capacitors, specifically designed for switch mode converter applications, are also available, and these are a better choice than general purpose devices. even better performance can be achieved with tantalum capacitors, although their cost is higher. very low values of esr can be achieved by using os-con capacitors (sanyo corporation, san diego, ca). these devices are fairly small, available with tape-and-reel packaging and have very low esr. the effects of capacitor selection on output ripple are demon- strated in figures 15, 16 and 17. these figures show the output of the same adp1111 converter that was evaluated with three different output capacitors. in each case, the peak switch current is 500 ma, and the capacitor value is 100 f. figure 15 shows a panasonic hf-series 16-volt radial cap. when the switch turns off, the output voltage jumps by about 90 mv and then decays as the inductor discharges into the capacitor. the rise in voltage indicates an esr of about 0.18 . in figure 16, the aluminum electrolytic has been replaced by a sprague 293d series, a 6 v tantalum device. in this case the output jumps about 30 mv, which indicates an esr of 0.06 . figure 17 shows an os-con 16Cvolt capacitor in the same circuit, and esr is only 0.02 . rev. a
adp1111 C9C rev. 0 figure 15. aluminum electrolytic figure 16. tantalum electrolytic figure 17. os-con capacitor if low output ripple is important, the user should consider the adp3000. because this device switches at 400 khz, lower peak current can be used. also, the higher switching frequency simplifies the design of the output filter. consult the adp3000 data sheet for additional details. diode selection in specifying a diode, consideration must be given to speed, forward voltage drop and reverse leakage current. when the adp1111 switch turns off, the diode must turn on rapidly if high efficiency is to be maintained. shottky rectifiers, as well as fast signal diodes such as the 1n4148, are appropriate. the forward voltage of the diode represents power that is not delivered to the load, so v f must also be minimized. again, schottky diodes are recommended. leakage current is especially important in low-current applications where the leakage can be a significant percentage of the total quiescent current. for most circuits, the 1n5818 is a suitable companion to the adp1111. this diode has a v f of 0.5 v at 1 a, 4 a to 10 a leakage, and fast turn-on and turn-off times. a surface mount version, the mbrs130t3, is also available. for switch currents of 100 ma or less, a shottky diode such as the bat85 provides a v f of 0.8 v at 100 ma and leakage less than 1 a. a similar device, the bat54, is available in a sot23 package. even lower leakage, in the 1 na to 5 na range, can be obtained with a 1n4148 signal diode. general purpose rectifiers, such as the 1n4001, are not suitable for adp1111 circuits. these devices, which have turn-on times of 10 s or more, are far too slow for switching power supply applications. using such a diode just to get started will result in wasted time and effort. even if an adp1111 circuit appears to function with a 1n4001, the resulting performance will not be indicative of the circuit performance when the correct diode is used. circuit operation, step-up (boost) mode in boost mode, the adp1111 produces an output voltage that is higher than the input voltage. for example, +12 v can be gener- ated from a +5 v logic power supply or +5 v can be derived from two alkaline cells (+3 v). figure 18 shows an adp1111 configured for step-up operation. the collector of the internal power switch is connected to the output side of the inductor, while the emitter is connected to gnd. when the switch turns on, pin sw1 is pulled near ground. this action forces a voltage across l1 equal to v in C v ce(sat) , and current begins to flow through l1. this current reaches a final value (ignoring second-order effects) of: i peak ? v in ? v ce ( sat ) l ? 7 s where 7 s is the adp1111 switchs on time. i lim v in sw1 fb gnd sw2 adp1111 5 4 + v in l1 d1 1n5818 c1 r2 r1 v out r3 (optional) 1 2 3 8 figure 18. step-up mode operation when the switch turns off, the magnetic field collapses. the polarity across the inductor changes, current begins to flow through d1 into the load, and the output voltage is driven above the input voltage. the output voltage is fed back to the adp1111 via resistors r1 and r2. when the voltage at pin fb falls below 1.25 v, sw1 turns on again, and the cycle repeats. the output voltage is therefore set by the formula: v out = 1. 2 5 v ? 1 + r 2 r 1 ? ? ? ? ? ? the circuit of figure 18 shows a direct current path from v in to v out , via the inductor and d1. therefore, the boost converter is not protected if the output is short circuited to ground. rev. a
adp1111 C10C rev. 0 circuit operation, step down (buck) mode) the adp1111s step down mode is used to produce an output voltage that is lower than the input voltage. for example, the output of four nicd cells (+4.8 v) can be converted to a +3 v logic supply. a typical configuration for step down operation of the adp1111 is shown in figure 19. in this case, the collector of the internal power switch is connected to v in and the emitter drives the inductor. when the switch turns on, sw2 is pulled up towards v in . this forces a voltage across l1 equal to v in C v ce C v out and causes current to flow in l1. this current reaches a final value of: i peak ? v in ? v ce ? v out l ? 7 s where 7 s is the adp1111 switchs on time. i lim v in sw1 sw2 4 gnd set ao adp1111 nc l1 d1 1n5818 r lim 100 1 + v in 2 3 6 7 5 nc c 2 + v out r2 r1 c l fb 8 figure 19. step-down mode operation when the switch turns off, the magnetic field collapses. the polarity across the inductor changes, and the switch side of the inductor is driven below ground. schottky diode d1 then turns on, and current flows into the load. notice that the absolute maximum rating for the adp1111s sw2 pin is 0.5 v below ground. to avoid exceeding this limit, d1 must be a schottky diode. if a silicon diode is used for d1, pin sw2 can go to C0.8 v, which will cause potentially damaging power dissipation within the adp1111. the output voltage of the buck regulator is fed back to the adp1111s fb pin by resistors r1 and r2. when the voltage at pin fb falls below 1.25 v, the internal power switch turns on again, and the cycle repeats. the output voltage is set by the formula: v out = 1. 2 5 v ? 1 + r 2 r 1 ? ? ? ? ? ? when operating the adp1111 in step-down mode, the output voltage is impressed across the internal power switchs emitter- base junction when the switch is off. to protect the switch, the output voltage should be limited to 6.2 v or less. if a higher output voltage is required, a schottky diode should be placed in series with sw2 as shown in figure 20. i lim v in sw1 sw2 fb gnd adp1111 l1 d1 r 3 1 + v in 2 3 5 8 4 c 2 + v out r2 r1 d2 c 1 d1, d2 = 1n5818 schottky diodes figure 20. step-down model, v out > 6.2 v if the input voltage to the adp1111 varies over a wide range, a current limiting resistor at pin 1 may be required. if a particular circuit requires high peak inductor current with minimum input supply voltage, the peak current may exceed the switch maxi- mum rating and/or saturate the inductor when the supply voltage is at the maximum value. see the limiting the switch current section of this data sheet for specific recommendations. increasing output current in the step-down regulator unlike the boost configuration, the adp1111s internal power switch is not saturated when operating in step-down mode. a conservative value for the voltage across the switch in step-down mode is 1.5 v. this results in high power dissipation within the adp1111 when high peak current is required. to increase the output current, an external pnp switch can be added (figure 21). in this circuit, the adp1111 provides base drive to q1 through r3, while r4 ensures that q1 turns off rapidly. because the adp1111s internal current limiting function will not work in this circuit, r5 is provided for this purpose. with the value shown, r5 limits current to 2 a. in addition to reducing power dissipation on the adp1111, this circuit also reduces the switch voltage. when selecting an inductor value for the circuit of figure 21, the switch voltage can be calculated from the formula: v = v + v 0.6 v + 0.4 v 1 v sw r5 q1(sat) ?? i lim v in sw1 sw2 fb gnd set ao adp1111 nc l1 d1 1n5821 r5 0.3 1 input 2 3 6 7 5 4 8 nc c input + r1 r2 c l + output r3 330 r4 220 q1 mje210 figure 21. high current step-down operation rev. a
adp1111 C11C rev. 0 positive-to-negative conversion the adp1111 can convert a positive input voltage to a negative output voltage as shown in figure 22. this circuit is essentially identical to the step-down application of figure 19, except that the output side of the inductor is connected to power ground. when the adp1111s internal power switch turns off, current flowing in the inductor forces the output (Cv out ) to a negative potential. the adp1111 will continue to turn the switch on until its fb pin is 1.25 v above its gnd pin, so the output voltage is determined by the formula: v out = 1. 2 5 v ? 1 + r 2 r 1 ? ? ? ? ? ? i lim v in sw1 sw2 fb gnd set ao adp1111 nc l1 d1 1n5818 r lim 1 input 2 3 6 7 5 4 8 nc c input + r1 r2 c l + output negative output figure 22. positive-to-negative converter the design criteria for the step-down application also apply to the positive-to-negative converter. the output voltage should be limited to |6.2 v| unless a diode is inserted in series with the sw2 pin (see figure 20.) also, d1 must again be a schottky diode to prevent excessive power dissipation in the adp1111. negative-to-positive conversion the circuit of figure 23 converts a negative input voltage to a positive output voltage. operation of this circuit configuration is similar to the step-up topology of figure 18, except the current through feedback resistor r2 is level-shifted below ground by a pnp transistor. the voltage across r2 is v out Cv beq1 . how- ever, diode d2 level-shifts the base of q1 about 0.6 v below ground thereby cancelling the v be of q1. the addition of d2 also reduces the circuits output voltage sensitivity to tempera- ture, which otherwise would be dominated by the C2 mv v be contribution of q1. the output voltage for this circuit is determined by the formula: v out = 1. 2 5 v ? r 2 r 1 unlike the positive step-up converter, the negative-to-positive converters output voltage can be either higher or lower than the input voltage. i lim v in sw1 sw2 fb gnd set ao adp1111 nc d1 1n5818 1 2 3 6 7 5 4 8 nc c2 + r1 10k c l + positive output r2 mje210 r lim negative input l1 d2 2n3906 q1 figure 23. adp1111 negative-to-positive converter limiting the switch current the adp1111s r lim pin permits the switch current to be limited with a single resistor. this current limiting action occurs on a pulse by pulse basis. this feature allows the input voltage to vary over a wide range without saturating the inductor or exceeding the maximum switch rating. for example, a particular design may require peak switch current of 800 ma with a 2.0 v input. if v in rises to 4 v, however, the switch current will exceed 1.6 a. the adp1111 limits switch current to 1.5 a and thereby protects the switch, but the output ripple will increase. selecting the proper resistor will limit the switch current to 800 ma, even if v in increases. the relationship between r lim and maximum switch current is shown in figure 6. the i lim feature is also valuable for controlling inductor current when the adp1111 goes into continuous-conduction mode. table i. component selection for typical converters input output output circuit inductor inductor capacitor voltage voltage current (ma) figure value part no. value notes 2 to 3.1 5 90 ma 4 15 h cd75-150k 33 f* 2 to 3.1 5 10 ma 4 47 h ctx50-1 10 f 2 to 3.1 12 30 ma 4 15 h cd75-150k 22 f 2 to 3.1 12 10 ma 4 47 h ctx50-1 10 f 5 12 90 ma 4 33 h cd75-330k 22 f 51230ma 447 h ctx50-1 15 f 6.5 to 11 5 50 ma 5 15 h47 f** 12 to 20 5 300 ma 5 56 h ctx50-4 47 f** 20 to 30 5 300 ma 5 120 h ctx100-4 47 f** 5C57ma 656 h ctx50-4 47 f 12 C5 250 ma 6 120 h ctx100-4 100 f** notes cd = sumida. ctx = coiltronics. * *add 47 from i lim to v in . **add 220 from i lim to v in . rev. a
adp1111 C12C rev. 0 this occurs in the step-up mode when the following condition is met: v out + v diode v in ? v sw < 1 1 ? dc where dc is the adp1111s duty cycle. when this relationship exists, the inductor current does not go all the way to zero during the time that the switch is off. when the switch turns on for the next cycle, the inductor current begins to ramp up from the residual level. if the switch on time remains constant, the inductor current will increase to a high level (see figure 24). this increases output ripple and can require a larger inductor and capacitor. by controlling switch current with the i lim resistor, output ripple current can be maintained at the design values. figure 25 illustrates the action of the i lim circuit. figure 24. figure 25. the internal structure of the i lim circuit is shown in figure 26. q1 is the adp1111s internal power switch that is paralleled by sense transistor q2. the relative sizes of q1 and q2 are scaled so that i q2 is 0.5% of i q1 . current flows to q2 through an internal 80 resistor and through the r lim resistor. these two resistors parallel the base-emitter junction of the oscillator- disable transistor, q3. when the voltage across r1 and r lim exceeds 0.6 v, q3 turns on and terminates the output pulse. if only the 80 internal resistor is used (i.e. the i lim pin is connected directly to v in ), the maximum switch current will be 1.5 a. figure 6 gives r lim values for lower current-limit values. 72khz osc v in power switch sw2 sw1 r lim driver 80 (internal) i lim i q1 200 v in (external) q2 adp1111 q1 q3 r1 figure 26. adp1111 current limit operation the delay through the current limiting circuit is approximately 1 s. if the switch on time is reduced to less than 3 s, accuracy of the current trip-point is reduced. attempting to program a switch on time of 1 s or less will produce spurious responses in the switch on time; however, the adp1111 will still provide a properly regulated output voltage. programming the gain block the gain block of the adp1111 can be used as a low-battery detector, error amplifier or linear post regulator. the gain block consists of an op amp with pnp inputs and an open-collector npn output. the inverting input is internally connected to the adp1111s 1.25 v reference, while the noninverting input is available at the set pin. the npn output transistor will sink about 300 a. figure 27a shows the gain block configured as a low-battery monitor. resistors r1 and r2 should be set to high values to reduce quiescent current, but not so high that bias current in the set input causes large errors. a value of 33 k for r2 is a good compromise. the value for r1 is then calculated from the formula: r 1 = v lobatt ? 1. 2 5 v 1. 2 5 v r 2 where v lobatt is the desired low battery trip point. since the gain block output is an open-collector npn, a pull-up resistor should be connected to the positive logic power supply. adp1111 1.25v ref gnd ao 5v r l 47k to processor r1 r2 v bat v in set 33k r1= ????????? v lb ?1.25v 35.1 a v lb = battery trip point figure 27a. setting the low battery detector trip point 200ma/div 200ma/div rev. a
adp1111 C13C rev. 0 the circuit of figure 27b may produce multiple pulses when approaching the trip point due to noise coupled into the set input. to prevent multiple interrupts to the digital logic, hysteresis can be added to the circuit (figure 27). resistor rhys, with a value of 1 m to 10 m , provides the hysteresis. the addition of rhys will change the trip point slightly, so the new value for r1 will be: r 1 = v lobatt ? 1. 2 5 v 1. 2 5 v r 2 ? ? ? ? ? ? ? v l ? 1. 2 5 v r l + r hys ? ? ? ? ? ? where v l is the logic power supply voltage, r l is the pull-up resistor, and r hys creates the hysteresis. adp1111 1.25v ref gnd ao 5v r l 47k to processor r1 r2 v bat v in set r hys 33k 1.6m figure 27b. application circuits all surface mount 3 v to 5 v step-up converter this is the most basic application (along with the basic step- down configuration to follow) of the adp1111. it takes full advantage of surface mount packaging for all the devices used in the design. the circuit can provide +5 v at 100 ma of output current and can be operated off of battery power for use in portable equipment. + l1 d1 c l 33 f output r3* (optional) mbrs120t3 20 h ctx20-4 (5v @ 100ma) input +3v i lim v in sw2 sw1 sense gnd set ao adp1111-5 nc 1 2 6 7 5 3 8 nc 4 figure 28. all surface mount +3 v to +5 v step-up converter 9 v to 5 v step-down converter this circuit uses a 9 v battery to generate a +5 v output. the circuit will work down to 6.5 v, supplying 50 ma at this lower limit. switch current is limited to around 500 ma by the 100 resistor. i lim v in sw1 sw2 sense gnd set ao adp1111-5 nc l1 c l 22 f d1 1n5818 r lim 100 1 + 15 h ctx15-4 input 2 3 6 7 5 4 8 nc output (9v in to 5v @ 150ma, 6.5v in to 5v @ 50ma) 9v figure 29. 9 v to 5 v step-down converter 20 v to 5 v step-down converter this circuit is similar to figure 29, except it supplies much higher output current and operates over a much wider range of input voltage. as in the previous examples, switch current is limited to 500 ma. i lim v in sw1 sw2 sense gnd set ao adp1111-5 nc l1 c l 47 f d1 1n5818 r lim 100 1 + 68 h ctx68-4 12v to 28v input 2 3 6 7 5 4 8 nc output (+5v @ 300ma) figure 30. 20 v to 5 v step-down converter +5 v to C5 v converter this circuit is essentially identical to figure 22, except it uses a fixed-output version of the adp1111 to simplify the design somewhat. i lim v in sw1 sw2 gnd set ao adp1111-5 nc l1 c l 33 f d1 1n5818 r lim 100 1 + 33 h ctx33-2 12v to 28v input 2 3 6 7 5 4 8 nc ?5v @ 75ma sense figure 31. +5 v to C5 v converter rev. a i lim v in sw1 sw2 gnd set ao adp1111-5 nc l1 c l 33 f d1 1n5818 r lim 100 1 + 33 h ctx33-2 5v to 25v input 2 3 6 7 5 4 8 nc ?5v @ 75ma sense
adp1111 C14C rev. 0 voltage-controlled positive-to-negative converter by including an op amp in the feedback path, a simple positive- to-negative converter can be made to give an output that is a linear multiple of a controlling voltage, vc. the op amp, an op196, rail-to-rail input and output amplifier, sums the currents from the output and controlling voltage and drives the fb pin either high or low, thereby controlling the on-board oscillator. the 0.22 resistor limits the short-circuit current to about 3 a and, along with the bat54 schottky diode, helps limit the peak switch current over varying input voltages. the external power switch features an active pull-up to speed up the turn-off time of the switch. although an irf9530 was used in the evaluation, almost any device that can handle at least 3 a of peak current at a vds of at least 50 v is suitable for use in this application, provided that adequate attention is paid to power dissipation. the circuit can deliver 2 w of output power with a +6-volt input from a control voltage range of 0 v to 5 v. i lim v in sw1 sw2 fb gnd set ao adp1111 nc 2 1 3 6 7 5 4 8 nc l1 c l 47 f 35v output r lim input + +5v to +12v 0.22 2k 3 4 6 7 2 1k 200k 39k v in 1n4148 irf9530 20 h d1 in5821 ?v out = ?5.13 *v c 2w maximum output 1n5231 ctx20-4 v c (0v to +5v) 2n3904 51 bat54 figure 32. voltage controlled positive-to-negative converter +3 v to C22 v lcd bias generator this circuit uses an adjustable-output version of the adp1111 to generate a +22.5 v reference output that is level-shifted to give an output of C22 v. if operation from a +5 volt supply is desired, change r1 to 47 ohms. the circuit will deliver 7 ma with a 3 volt supply and 40 ma with a 5 volt supply. l1 c l 0.1 f output 25 h d1 1n4148 1n5818 1n5818 732k 42.2k 22 f 4.7 f +3v 2xaa cells r lim 100 ?22v output 7ma @ 2v input l1 = ctx25-4 i lim v in sw1 sw2 fb gnd set ao adp1111 nc 1 2 3 6 7 5 4 8 nc + + figure 33. 3 v to C22 v lcd bias generator high power, low quiescent current step-down converter by making use of the fact that the feedback pin directly controls the internal oscillator, this circuit achieves a shutdown-like state by forcing the feedback pin above the 1.25 v comparator threshold. the logic level at the 1n4148 diode anode needs to be at least 2 v for reliable standby operation. the external switch driver circuit features an active pull-up device, a 2n3904 transistor, to ensure that the power mosfet turns off quickly. almost any power mosfet will do as the switch as long as the device can withstand the 18 volt v gs and is reasonably robust. the 0.22 resistor limits the short-circuit current to about 3 a and, along with the bat54 schottky diode, helps to limit the peak switch current over varying input voltages. c l 220 f + irf9540 d1 in5821 nc r lim input +8v to +18v 0.22 2k 1n4148 bat54 2n3904 g s d 121k 40.2k 51 1n4148 +5v 500ma operate /standby 2v v in 5 i lim v in sw1 sw2 fb gnd set ao adp1111 1 2 3 6 7 5 4 8 nc li 20 h li = coiltronics ctx20-4 figure 34. high power, low quiescent current step-down converter notes 1. all inductors referenced are coiltronics ctx-series except where noted. 2. if the source of power is more than an inch or so from the converter, the input to the converter should be bypassed with approximately 10 f of capacitance. this capacitor should be a good quality tantalum or aluminum electrolytic. rev. a
adp1111 rev. ? a \ 15\ outline dimensions compliant to jedec standards ms-001 controlling dimensions are in inches; millimeter dimensions (in parentheses) are rounded-off inch equivalents for reference only and are not appropriate for use in design. corner leads may be configured as whole or half leads. 070606-a 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) seating plane 0.015 (0.38) min 0.210 (5.33) max 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) 0.070 (1.78) 0.060 (1.52) 0.045 (1.14) 8 1 4 5 0.280 (7.11) 0.250 (6.35) 0.240 (6.10) 0.100 (2.54) bsc 0.400 (10.16) 0.365 (9.27) 0.355 (9.02) 0.060 (1.52) max 0.430 (10.92) max 0.014 (0.36) 0.010 (0.25) 0.008 (0.20) 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.195 (4.95) 0.130 (3.30) 0.115 (2.92) 0.015 (0.38) gauge plane 0.005 (0.13) min figure 35. 8-lead plastic dual in-line package (pdip) narrow body (n-8) dimensions shown in inches and (millimeters) controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design. compliant to jedec standards ms-012-a a 012407-a 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) 0.50 (0.0196) 0.25 (0.0099) 45 8 0 1.75 (0.0688) 1.35 (0.0532) seating plane 0.25 (0.0098) 0.10 (0.0040) 4 1 85 5.00 (0.1968) 4.80 (0.1890) 4.00 (0.1574) 3.80 (0.1497) 1.27 (0.0500) bsc 6.20 (0.2441) 5.80 (0.2284) 0.51 (0.0201) 0.31 (0.0122) coplanarity 0.10 figure 36. 8-lead standard small outline package (soic_n) narrow body (r-8) dimensions shown in millimeters and (inches) ordering guide model 1 output voltage temperature range package description package option adp1111anz-12 12 v ?40c to +85c 8-lead plastic dual in-line package [pdip] n-8 adp1111anz-3.3 3.3 v ?40c to +85c 8-lead plastic dual in-line package [pdip] n-8 ADP1111ANZ-5 5 v ?40c to +85c 8-lead plastic dual in-line package [pdip] n-8 adp1111arz adj ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-12 12 v ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-12-reel 12 v ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-3.3 3.3 v ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-5 5 v ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-5-reel 5 v ?40c to +85c 8-lead standard small outline package [soic_n] r-8 adp1111arz-reel adj ?40c to +85c 8-lead standard small outline package [soic_n] r-8 1 z = rohs compliant part. ?1996C2009 analog devices, inc. all rights reserved. trademarks and registered trademarks are the prop erty of their respective owners. d08365-0-11/09(a)

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